IEEE JOURNAL OF SOLID-STATE CIRCUITS, VOL. 35, NO. 12, DECEMBER 2000
1877
A CMOS Nested-Chopper Instrumentation Amplifier
with 100-nV Offset
Anton Bakker, Kevin Thiele, and Johan H. Huijsing, Fellow, IEEE
Abstract—A CMOS nested-chopper instrumentation amplifier
is presented with a typical offset of 100 nV. This performance is obtained by nesting an additional low-frequency chopper pair around
a conventional chopper amplifier. The inner chopper pair removes
the 1
noise, while the outer chopper pair reduces the residual
offset. The test chip is free from 1 noise and has a thermal noise
of 27 nV/ Hz consuming a total supply current of 200 A.
Index Terms—Chopper amplifiers, instrumentation amplifiers,
low-offset amplifiers, offset cancellation techniques.
I. INTRODUCTION
I
N many applications, such as sensor interfaces, the overall
performance of the system is limited by the offset and noise
of the input amplifiers. This problem has been growing in the
past years, because of the shift from bipolar to CMOS pronoise and offset.
cesses which have significantly higher
Also a conventional offset-cancellation technique such as trimming, which is widely used in bipolar technology, is much less
beneficial in CMOS technology because it can not reduce the
noise. Solutions that can remove both offset and
noise
are found in the dynamic offset-cancellation techniques. Examples of these are the autozero and chopper techniques, which
will be explained in this paper. Derivatives of these techniques
are found in all commercial ultra-low-offset CMOS operational
amplifiers. Typical offset figures of these kinds of operational
amplifiers are in the range of 10 V. Although these offset figures are already very low, still some applications require an even
lower offset.
In this paper, an instrumentation amplifier for read-out of a
spinning-current Hall plate is described [1]. This Hall plate is
integrated in a standard CMOS process and has an offset of less
than 500 nV. The offset of the instrumentation amplifier needs
therefore to be well below this 500 nV. This paper discusses
the design and realization of this ultra-low-offset instrumentation amplifier. A new dynamic offset-cancellation technique is
shown that can reduce the offset of a CMOS amplifier to typically 100 nV.
Manuscript received April 17, 2000; revised July 15, 2000.
A. Bakker and K. Thiele are with Philips Semiconductors, Sunnyvale, CA
94088 USA.
J. H. Huijsing is with the Electronic Instrumentation Laboratory, Delft Institute of MicroElectronics and Submicron Technology (DIMES), Delft University
of Technology, The Netherlands.
Publisher Item Identifier S 0018-9200(00)10050-2.
Fig. 1.
Noise power spectrum of standard CMOS operational amplifier.
II. DYNAMIC OFFSET-CANCELLATION TECHNIQUES
A. Offset and Noise in CMOS Amplifiers
A conventional CMOS amplifier has a typical input-referred
noise spectrum, as shown in Fig. 1. For rather high frequencies, the noise can be considered as frequency independent or
white. This is usually called the thermal noise floor. At low frequencies, the noise power is increasing almost linearly with denoise.
creasing frequency and is therefore commonly called
noise becomes dominant over
The frequency at which the
noise corner frequency
.
the white noise is called the
At very low frequencies, offset becomes the dominant error.
Although offset is usually modeled as a time-invariant voltage
source, it may change due to aging and temperature variations.
This implies that it has a certain bandwidth and can therefore be
considered as a very low-frequency noise source.
B. Classification
Due to historical reasons, some confusion has arisen in the
naming conventions of the different dynamic offset-cancellation
techniques. Nowadays, it is generally accepted to distinguish
two main groups: autozeroing and chopping [2]–[4]. The fundamental difference between them is the offset handling. While
the autozero principle first measures the offset and subtracts it in
a next phase, chopping modulates the offset to higher frequencies, which will be explained further in this paper.
In data books and literature, many derivatives of these
two basic offset-cancellation techniques can be found, like
correlated double-sampling [2], [4], ping-pong opamps [5], [6],
self-calibrating opamps [7], synchronous detection, the twoor three-signal approach [8], and dynamic element matching
(DEM). The name chopper stabilization is even used for both
autozeroing and chopping techniques [2], [9].
Table I shows the above-mentioned techniques classified into
the two main groups: autozeroing and chopping. The next para-
0018–9200/00$10.00 © 2000 IEEE
1878
IEEE JOURNAL OF SOLID-STATE CIRCUITS, VOL. 35, NO. 12, DECEMBER 2000
TABLE I
CLASSIFICATION OF DYNAMIC OFFSET-CANCELLATION TECHNIQUES
Fig. 3. Noise power spectrum of autozeroed amplifier.
Fig. 2. Principle of autozero technique.
graphs will discuss the basic characteristics of the autozero and
the chopper technique.
C. Autozero Technique
The principle of the autozero technique is shown in Fig. 2.
The main characteristic of the autozero technique and its derivatives is that the offset cancellation is done in two phases. A samwhen the offset is measured and sampled on
pling phase
and an amplification phase
when the sampled offset is subtracted from the input signal and amplified. This technique is
very well-known and an improved derivative is applied in all
commercially available chopper-stabilized opamps. The reason
to mention it here is the noise performance of this technique.
Besides the offset, the autozero technique also removes the
noise of the amplifier, which makes sense because offset
noise. To remove all the
can be considered as low-frequency
noise the autozeroing frequency
should be higher than
noise corner frequency. The typical noise power specthe
trum of an autozero amplifier is shown in Fig. 3. The residual
noise at frequencies lower than is almost white. However, it is
not equal to the thermal noise floor, as it is increased by the ratio
of the unity-gain bandwidth of the amplifier and the autozeroing frequency . The reason for this is that due to the sampling action, high-frequency components are folded back to the
baseband. The higher the bandwidth of the amplifier, the more
. A more precise explananoise is sampled on the capacitor
tion is given by Enz et al. in [2]. In practical situations, the ratio
is somewhere between three and five. This is the fundamental reason why, for example, chopper-stabilized opamps always have rather high noise figures in the order of 70 nV/ Hz.
D. Chopper Technique
The principle of the chopper technique is shown in Fig. 4.
is modulated to the chopping frequency,
The input signal
amplified and modulated back to the baseband. The offset is
modulated only once and appears at the chopping frequency and
its odd harmonics. These frequency components need to be removed by a low-pass filter. Next to the frequency domain, the
chopping principle can also be explained in the time-domain.
is periodically inverted by the
In that case, the input signal
first multiplier or chopper. After amplification, the inverted and
amplified signal is inverted for the second time, resulting again
in a dc signal. The offset is periodically inverted only once and
therefore appears as a square wave at the output.
In contrast to the increased white noise component of autozero amplifiers, the baseband noise of chopper amplifiers is
almost equal to the wideband thermal noise, assuming again
noise corner
that the chopping frequency is higher than the
frequency. The typical noise power spectrum of a chopper amnoise components
plifier is shown in Fig. 5. The folded
are omitted for simplicity. The reason why the residual noise of
a chopper amplifier is fundamentally lower than that of an autozero amplifier is that the input signal of a chopper amplifier is
not sampled, which makes it impossible for wideband thermal
noise to fold back into the baseband.
The lower noise of the chopper technique is the main reason
to use this technique for read-out of our spinning-current Hall
plate. However, the residual offset of the chopper technique is
still too high for this application. The rest of this paper will
therefore focus on techniques to further reduce this residual
offset of a chopper amplifier.
E. Origins of Offset in Chopper Amplifiers
The residual offset of a chopper amplifier is in the order of a
few tens of V. To be able to reduce this offset, first the origins
of the offset need to be explored.
The residual offset of a chopper amplifier originates mainly
from the spikes of the input chopper [2]. In turn, these spikes
originate from the charge injection mismatch of the switches.
After demodulation of the spikes a residual offset occurs. This
is schematically shown in Fig. 6. It can be seen that the residual
is determined by the number of the spikes and the
offset
energy content of the spikes. This energy content is dictated
and the mismatch in parasitic caby the input impedance
and the
pacitive coupling between the chopping signal
. There are three main
input lines, which is presented by
options to reduce the residual offset: 1) lowering the chopping
frequency; 2) lowering the input impedance; or 3) lowering the
charge injection. However, lowering the chopping frequency is
not a real solution, because the chopping frequency should be
BAKKER et al.: CMOS NESTED-CHOPPER INSTRUMENTATION AMPLIFIER
Fig. 4.
1879
Chopping principle including signals in frequency and time domain.
Fig. 5. Noise power spectrum of chopper amplifier.
higher than the
noise corner frequency to remove the
noise. The input resistance
is dictated by the input signal
source and can usually not be lowered by the designer. Charge
injection is mainly dictated by the process choice and can be
minimized by the designer by using small transistors that are
well matched. The final conclusion is that except for careful
layout design, the designer can not improve the residual offset
in a straightforward manner.
F. Techniques to Reduce the Residual Offset
An interesting nonstraightforward method to reduce the
residual offset is shown by Menolfi et al. [10]. They remark
that the energy content of the spikes is mainly located at higher
harmonics of the chopping frequency, while the energy of the
modulated signal is mainly located at the fundamental of the
chopping frequency. If the modulated signal that includes the
spikes is low-pass or bandpass filtered, almost all spikes are
removed, while only a small part of the signal is lost. This
idea is schematically shown in Fig. 7. Compared to a conventional chopper amplifier, a bandpass filter is added within the
of this bandpass filter is
amplifier. The center frequency
. The quality factor
equal to the chopping frequency
should be high, to increase the attenuation of the unwanted
Fig. 6. Residual offset caused by spikes. (a) Spike signal. (b) Demodulation
signal. (c) Demodulated spike.
spikes. However, a high introduces gain accuracies if there
and . For a mismatch between
is a mismatch between
and
of 1%, a good compromise value for is between
three and five. This value gives a residual offset of 500 nV
which is the best reported value so far. As already mentioned
before, the major drawback of this circuit is the gain accuracy.
This gain accuracy is dependent on the quality factor and the
and . This implies that already in a
matching between
relatively small temperature range, large deviations in accuracy
will occur. The accuracy can also not be improved by applying
feedback because of stability problems caused by the phase
response of the bandpass filter. In conclusion, this technique
significantly reduces the residual offset at the cost of reduced
gain accuracy.
1880
Fig. 7.
IEEE JOURNAL OF SOLID-STATE CIRCUITS, VOL. 35, NO. 12, DECEMBER 2000
Chopper amplifier with bandpass filter to improve residual offset.
Fig. 8. Nested chopper amplifier principle.
III. NESSTED-CHOPPER TECHNIQUE
A new technique to reduce the residual offset of a chopper
amplifier is shown in Fig. 8, which will be referred to as the
nested-chopper technique. The basic idea is to consider a conventional chopper amplifier as a regular amplifier without
noise and a reduced offset. The offset of this amplifier can be reduced by applying another pair of choppers, but now operating
at a much lower frequency. This frequency can be much lower
noise corner frequency, because the
noise is althan the
ready removed by the inner pair of choppers. Because the outer
pair is working at a much lower frequency the residual offset due
to spikes of these choppers is much lower. The corresponding
signals of the nested-chopper amplifier are shown in Fig. 9. The
phase relationship of both signals is not important. This implies
in practical situations that both signals will have synchronous
edges. The spikes that are generated by the high chopping freare modulated by the output chopper with a
quency
. The average energy of the spikes has befrequency
come zero now, resulting in a residual offset that is theoretically zero. If we take into account the spikes generated by the
low-frequency choppers, the theoretically achievable improveand
. For
ment in residual offset is the ratio of
and
of 2 kHz and 20 Hz,
practical values of
respectively, the residual offset will be 100 times less. This implies that it should be possible to reduce the residual offset to
less than 100 nV. Compared to the above-mentioned bandpass
chopper amplifier, the nested-chopper amplifier can be fed back,
which implies that the gain-accuracy can be very high over a
wide temperature range. Also the implementation is very simple
as only one extra pair of choppers and a clock signal are needed.
A disadvantage is that the maximum input signal frequency is
. However, a bandwidth of a few tens of
reduced to half
Hertz is sufficient for many data-acquisition applications.
The conclusion on the proposed nested-chopper amplifier
technique is that the offset can be reduced to values as low as
Fig. 9. Reduction of residual offset by nested chopper amplifier. (a) Spikes
after first demodulator. (b) Low-frequency modulation signal. (c) Spikes after
second demodulator.
100 nV, without increasing the noise and with the possibility to
apply feedback.
IV. REALIZATION
To test and prove this theory, an instrumentation amplifier
employing the proposed nested-chopper technique is designed. The schematic is shown in Fig. 10. A major problem
in testing low-offset amplifiers is the effect of parasitic
thermocouples at the input when connecting an external test
signal. To avoid disturbances by these thermocouple effects,
we tested the system with an on-chip spinning-current
magnetic Hall sensor [1]. Switching off the Hall plate’s bias
current makes the signal at the input of the amplifier zero
without changing the source impedance. The amplification
,
,
factor of the amplifier is determined by resistors
and is set to 100.
and
BAKKER et al.: CMOS NESTED-CHOPPER INSTRUMENTATION AMPLIFIER
1881
Fig. 10.
Nested chopper instrumentation amplifier with spinning-current Hall sensor input.
Fig. 11.
Detailed schematic of chopper opamp including feedback resistors.
The low-pass filter is a first-order one with a 3-dB roll-off
frequency of 3 Hz. This low-pass filter is added externally, because of the required large RC constant. In a commercial version, this low-pass filter can be integrated on the chip by making
it a part of an integrating A/D converter, such as a sigma–delta
A/D converter.
The instrumentation amplifier can be divided in two equal
amplifiers. One of these amplifiers is shown in detail in Fig. 11.
The amplifier is a four-stage operational amplifier. The first
, and
is a low-gain
stage which is formed by
low-noise stage. This stage has a gain of approximately twenty.
The noise performance is optimized firstly by using resistors
and
instead of an active load and secondly by choosing
and
in such a
a high over ratio of input transistors
way that they are biased in weak inversion. The first stage deter-
mines the overall noise performance and has an equivalent input
thermal noise of approximately 15 nV/ Hz while drawing only
,
35- A tail current. The second stage consists of
and is optimized for high gain and low transconducand
tance. The high gain is necessary to reduce the influence of offsets of the subsequent stages outside the choppers. The reason
for the low transconductance is to have a low unity-gain frequency which implies an intrinsic filtering of the modulated
offset. This will be discussed more elaborately later. The low
transconductance is achieved by using a small tail current of
only 500 nA and a very low W/L ratio. The first two stages of the
amplifier are kept fully differential to reduce effects of charge
injection of the chopper switches. To define the common-mode
and
, a common-mode control
voltage on the drains of
circuit is necessary. This is done by measuring the common-
1882
IEEE JOURNAL OF SOLID-STATE CIRCUITS, VOL. 35, NO. 12, DECEMBER 2000
Fig. 13. Input referred offset versus chophigh frequency; resolution of
measurement is 50 nV.
Fig. 12.
Microphotograph of the test chip.
TABLE II
PERFORMANCE SUMMARY
mode current through
and
, and use this value to
and
. However, in the figure this is
control the currents
omitted for simplicity.
, and
and
The third stage is formed by
acts as a Miller stage. This stage splits the dominant poles of the
high-gain second and third stages. The fourth stage is formed by
and
and has a gain of approximately one.
One of the reasons for the Miller stage is to assure stability
of the opamp. However, this stage also acts as a low-pass filter
for the modulated offset of the first two stages. To have the
most aggressive filtering, the transconductance of the first two
stages should be as low as possible. However, the first stage
should have a high transconductance to achieve low noise
performance, which implies a trade-off. The second stage has
therefore a transconductance that is made as low as possible
without being a dominant noise source for the whole amplifier.
With a gain of 20 of the first amplifier, the transconsuctance of
the second amplifer can be 400 times lower to still not be the
dominant noise source. For our circuit, the transconductance
is set to 700 A/V and for the second
of the first stage,
equals 2 A/V. The unity-gain frequency is given by
stage,
, which equals 400 kHz. For an amplication
set by R41 and R42 ton100, this gives an amplifier bandwidth
of 4 kHz. The open-loop dc gain is more than 130 dB.
The current consumption of the instrumentation amplifier
is 200 A. A chip microphotograph is shown in Fig. 12. The
circuit is implemented in a single-poly double-metal 1.6- m
CMOS process. The die area is 6 mm . A large part of the die
area is occupied by the two 16-pF metal1–metal2 capacitors.
These capacitors could not be made out of MOS capacitors,
because they need to be very linear, because of the switching.
Because our process lacks the availability of double-poly linear
capacitors, metal1–metal2 capacitors were the only possible
implementation.
The higher chopping frequency is applied externally. The
is derived from
lower chopping frequency
by a frequency divider of 128. The divider value can also be
changed to 512 for test purposes.
V. MEASUREMENT RESULTS
Nine chips from two different batches have been tested. The
has been measured for
input referred offset versus
between 2 and 50 kHz. Experimental results
values of
is 2 kHz and a corresponding
show that for
of 16 Hz, the input referred offset is below 100 nV for all nine
samples. It also shows a significant increase for higher frequenbetween 2 and 8 kHz are
cies. The results for values of
shown in Fig. 13. The minimal value of 2 kHz is exactly equal
noise corner frequency.
to the
A remarkable result is that the offset is not dependent on
, but on
. The reason for this seems to be the
mismatch of the on-resistance of the low-frequency chopper
switches. This is explained in Fig. 14. If the input impedances
are not exactly equal, the area under the spikes do not completely cancel. This results in a residual offset due to the highfrequency spikes that is not exactly zero. If this is true, the dominant contribution to the residual offset does not come from the
low-frequency choppers but from the high-frequency ones. This
explains why the residual has approximately the value that could
.
be expected from theory, but is still dependent on
The noise is measured in a bandwidth of 0.1–3 Hz and was
found to be 27 nV/ Hz, which is in very good concordance with
the simulated thermal noise. The common-mode rejection ratio
is also measured in a 0.1–3-Hz band and was found to be over
BAKKER et al.: CMOS NESTED-CHOPPER INSTRUMENTATION AMPLIFIER
Fig. 14.
Explanation of the result that the residual offset is dependent on f
140 dB. This shows again the excellent common-mode rejection ratio (CMRR) performance of chopper amplifiers, which
was already known from previous designs [2]. A performance
summary is shown in Table II.
VI. CONCLUSION
In this paper, a nested-chopper technique is presented and
compared with other dynamic offset-cancellation techniques. It
is shown that this new technique can significantly reduce the
offset of conventional chopper amplifiers at the cost of only one
additional chopper pair. A nested-chopper instrumentation amplifier for a spinning-current Hall plate is realized and measurement results show a residual offset of only 100 nV. This offset
value is the lowest ever reported. A disadvantage is the limita.
tion of the maximum input signal frequency to half
However, a bandwidth of a few tens of hertz is sufficient for
many data-acquisition applications.
1883
.
[10] C. Menolfi and Q. Huang, “A fully integrated CMOS instrumentation
amplifier with submicrovolt offset,” IEEE J. Solid-State Circuits, vol.
34, pp. 415–420, Mar. 1999.
Anton Bakker was born in Amsterdam, The Netherlands, on July 11, 1968. In 1991, he received the
M.Sc. degree in electrical engineering from the Delft
University of Technology, Delft, The Netherlands.
In 1996, he started his Ph.D. project on CMOS smart
temperature sensors. During this research period, he
designed a number of temperature sensors for Philips
Semiconductors, Sunnyvale, CA. He received the
Ph.D. degree from Delft University in 2000.
In 1991, he joined the Werkgroep Elektrotechnisch
Practicum, where he developed a laboratory course
for second-year students on the design of complex integrated circuits. In 1993,
he joined the Electronic Instrumentation Laboratory where he was involved in
a European Project (ESPRIT) on the design of an ultra-low-power tempersature
sensor. During this project, he spent three months at CSEM, Neuchâtel, Switzerland, to implement his design. He is currently with Philips Semiconductors.
REFERENCES
[1] A. Bakker, A. A. Bellekom, S. Middelhoek, and J. H. Huijsing,
“Low-offset low-noise 3.5-mW CMOS spinning-current Hall effect
sensor with integrated chopper amplifier,” in Proc. Eurosensors XIII,
Sept. 1999, pp. 1045–1048.
[2] C. C. Enz and G. C. Temes, “Circuit techniques for reducing the effects
of opamp imperfections: Autozeroing, correlated double sampling, and
chopper stabilization,” Proc. IEEE, vol. 84, pp. 1584–1614, Nov. 1996.
[3] C. C. Enz, E. A. Vittoz, and F. Krummenacher, “A CMOS chopper amplifier,” IEEE J. Solid-State Circuits, vol. SC-22, pp. 335–342, June
1987.
[4] K. C. Hsieh, P. R. Gray, D. Senderowicz, and D. G. Messerschmitt,
“A low-noise chopper stabilized differential switched-capacitor filtering
technique,” IEEE J. Solid-State Circuits, vol. SC-16, pp. 708–715, Dec.
1981.
[5] C. G. Yu and R. L. Geiger, “An automatic offset compensation scheme
with ping-pong control for CMOS operational amplifiers,” IEEE J.
Solid-State Circuits, vol. 29, pp. 601–610, May 1994.
[6] I. E. Opris and G. T. A. Kovacs, “A rail-to-rail ping-pong opamp,” IEEE
J. Solid-State Circuits, vol. 31, pp. 1320–1324, Sept. 1996.
[7] (1999, May) TLC4501, Self-calibrating operational amplifier. Texas Instruments Inc., Dallas, TX. [Online]. Available: http://www.ti.com
[8] G. C. M. Meijer, “Concepts and focus point for intelligent sensor systems,” Sensors and Actuators, vol. 41, pp. 183–191, 1994.
[9] (1999, June) LTC1050, Precision chopper stabilized operational amplifier with internal capacitors. Linear Technology, Milpitas, CA. [Online].
Available: http://www.linear.com
Kevin Thiele, biography and photograph not available at time of publication.
Johan H. Huijsing (SM’81–F’97) was born in Bandung, Indonesia, on May 21, 1938. He received the
M.Sc. degree in electrical engineering from the Delft
University of Technology, Delft, The Netherlands, in
1969, and the Ph.D. degree from the same university
in 1981.
He has been an Assistant and Associate Professor
in electronic instrumentation with the Faculty
of Electrical Engineering, Delft University of
Technology, since 1969, where he became a full
Professor in the Chair of electronic instrumentation
in 1990. From 1982 through 1983, he was a Senior Scientist at Philips Research
Labs, Sunnyvale, CA. Since 1983, he has been a Consultant for Philips. His
research work is particulary focused on the systematic analysis and design of
operational amplifiers and integrated smart sensors. He is author or co-author
of some 150 scientific papers, 20 patents and four books, and co-editor of
five books. He is initiator and Co-Chairman of the International Workshop on
Advances in Analog Circuit Design, which has been held annually since 1992.
He is Chairman of the biannual National Workshop on Sensor Technology,
since 1991, and Chairman of the Dutch STW Platform on Sensor Technology.