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Novel Defected Ground Structure and Two-Side
Loading Scheme for Miniaturized Dual-Band
SIW Bandpass Filter Designs
Shanshan Xu, Kaixue Ma, Senior Member, IEEE, Fanyi Meng, Student Member, IEEE, and
Kiat Seng Yeo, Senior Member, IEEE
Abstract—This letter reports a novel back-to-back E-shaped
defected ground structure (DGS) and two-side loading scheme for
miniaturized dual-band substrate integrated waveguide (SIW)
bandpass ﬁlter (BPF) designs. The SIW loaded by novel DGS
supports evanescent-mode wave propagation below the cut-off
frequency of SIW, while achieves two transmission poles at the
passband and two controllable transmission zero points. Furthermore, the novel loading scheme by loading different-sized DGS
resonators on two sides of SIW is proposed for dual-band BPF
designs. To validate the design and analysis, a 2.4/5.2 GHz ﬁlter
prototype was designed. In good agreement with theoretical results, the measured results of the dual-band ﬁlter achieve insertion
losses of 3.6 and 3.1 dB, and fractional bandwidths of 5.8% and
6.45% at 2.4 and 5.2 GHz, respectively. The ﬁlter size is only
420 mil.
408 mil
Index Terms—Back-to-back E-shaped, bandpass ﬁlter (BPF),
complementary open-loop resonator (COLR), complementary
split ring resonator (CSRR), defected ground structure (DGS),
dual-band ﬁlters, miniaturization, substrate integrated waveguide
(SIW).
I. INTRODUCTION
S
UBSTRATE integrated waveguide (SIW), synthesized on
planar substrate using periodic arrays of metalized vias, has
been widely used as the platform for implementing low-cost and
integrated waveguide ﬁlters [1]. With the merits of low-loss and
high power handling capability, ﬁlters based on SIW technology
still suffer from bulky sizes compared to their microstrip counterparts [2].
To circumvent with this issue, the bandpass ﬁlters (BPFs)
with forward-wave or evanescent-mode wave which can propagate below SIW cut-off frequency
by loading DGS on SIW
sides, were demonstrated in [3]–[8] for ﬁlter miniaturization. In
Fig. 1. Conﬁguration and equivalent circuit of (a) the complementary
open-loop resonator [10] and (b) the proposed back-to-back E-shaped DGS
unit. (Grey zone: metallization; White zone: defected area).
these works, the primary investigation was leveraged on complementary split ring resonator (CSRR) [9] and complementary
open-loop resonator (COLR) as in Fig. 1(a). The dual-mode
and dual-band operations were demonstrated in [5] and [6], by
means of loading two same- or different-sized CSRRs on the top
side of SIW. To obtain lower resonant frequencies, same-sized
CSRRs/COLRs were loaded on two sides of SIW in [3] and [4].
In this letter, a novel DGS shape named as back-to-back
E-shaped DGS is proposed as shown in Fig. 1(b). By loading
the back-to-back E-shaped DGS on either top or bottom sides
of SIW, a dual-mode frequency response with two transmission
zero points (ZPs) is obtained. To obtain dual-band response, a
new two-side loading scheme with different-sized resonators on
top and bottom sides is also proposed in this letter. To validate
the concept, a dual-band (2.4/5.2 GHz) BPF is designed using
Rogers RO4003 (
,
) with thickness of
8 mil. The fabricated ﬁlter achieves good skirt selectivity, high
inter-band isolation, good spurious response, and compact size.
II. BACK-TO-BACK E-SHAPED DGS
A. Back-to-Back E-Shaped DGS
As shown in Fig. 1(b), the back-to-back E-shaped DGS is
formed by two E-shapes in back-to-back orientation. From its
physical conﬁguration, the E-shaped DGSs can be modeled as
Fig. 3. EM investigation of even, odd frequencies and coupling coefﬁcient
versus parameter . (Parameters of structure in Fig. 1 given in caption of Fig. 2).
. In Fig. 2(a), the simulation results of the circuit model are
well agreed with EM simulation results. Even- and odd-mode
analysis is adopted to study the resonant properties. Using circuits in Fig. 2(c), the odd-/even-mode input admittances are derived as in (1) and (2)
(1)
Fig. 2. Investigation of SIW loaded by back-to-back E-shaped DGS and COLR
on the top side: (a) conﬁgurations and transmission responses; (b) equivalent
circuit model of SIW loaded by unit back-to-back E-shaped DGS; (c) evenand odd-mode circuits. (Grey zone: metallization; White zone: defected area;
Physical parameters: for SIW
,
,
,
, for
single ring CSRR
,
,
, for back-to-back E-shaped
DGS:
,
,
,
,
,
with unit in mil;
Circuit elements:
,
,
,
,
,
, and
).
resonators ( and ) with
and
denoting their mutual magnetic and electric couplings. The resonant frequencies
of each E-shaped DGS are determined by , , , , and .
(2)
The odd- and even-mode transmission poles are located when
and
approaches inﬁnity
(3)
(4)
B. Transmission Responses
The transmission responses are investigated by using
ANSYS's HFSS V14 software packages. As shown in Fig. 2(a),
the unloaded SIW has a high-pass frequency response with
at 9.5 GHz.
First, the SIW is loaded by COLR unit as the reference. It is
observed that a transmission pole is created at 6 GHz, with one
ZP at 6.9 GHz.
Secondly, the SIW is loaded by the proposed DGS in the
same size. Compared to COLR, the proposed DGS achieves
two transmission poles at 5.3 and 5.35 GHz as shown in Fig.
2(a), that potential beneﬁts circuit miniaturization, enhances frequency selectivity by generating two transmission ZPs, and extends upper stopband attenuation of 20 dB to 9.5 GHz.
As shown in Fig. 2(b), the equivalent circuit models are developed for the proposed ﬁlter. The two sides of SIW are treated
as a two-wire transmission lines while the via arrays are modeled as inductors
[5]. The input coupling comprises of both
magnetic and electric couplings that are represented as
and
is inﬂuenced by
where only odd-mode resonant frequency
the mutual coupling elements
and
. It is investigated by
using EM simulation as the results given in Fig. 3 by varying
parameter and extracting the coupling coefﬁcients
as
(5)
It is veriﬁed that
is much more sensitive to the variation of
while
has negligible frequency shift. On the other hand,
the transmission ZPs are determined using
[11]
(6)
which are controlled by the input and mutual couplings.
Based on the EM extraction parameters, the single band
second-order ﬁlter is designed. With a targeted FBW, the
and the coupling coefﬁcient
external quality factor
between two DGS resonators are determined through the circuit
Fig. 4. A second-order BPF design based on SIW loaded by back-to-back
E-shaped DGS unit on the bottom side. (Physical parameters: same as the
caption of Fig. 2 with
; Circuit elements:
,
,
,
,
,
, and
).
DGS is much smaller than that of the 2.4 GHz one. As a result,
5.2 GHz ﬁlter has relatively less effect on the 2.4 GHz ﬁlter.
The dual-band ﬁlter can be designed by EM optimization of the
stacked 5.2 GHz ﬁlter on top and 2.4 GHz on bottom. Thus,
compared to parallel loading two different-sized resonators on
the same side [6], the size of proposed dual-band ﬁlter can be
reduced by half.
Fig. 6 shows the fabricated dual-band ﬁlter with a compact
size of 408 mil 420 mil (i.e.,
,
is the
2.4 GHz guided wavelength). The measured results agree with
the EM simulation result. For the 2.4 GHz band, the insertion
loss is 3.6 dB with a 3 dB FBW of 5.8%. For the 5.2 GHz band,
the insertion loss is 3.1 dB with FBW of 6.45%. The return
losses are better than 15 dB in both passbands. The designed
dual-band ﬁlter shows a sharp roll-off and high inter-band isolation better than 34 dB due to the generation of two transmission ZPs. The upper stopband with more than 20 dB attenuation
extends to 12 GHz.
IV. CONCLUSION
Fig. 5. The 2.4/5.2 GHz dual-band SIW BPF. (Physical parameters: Top side
,
,
,
,
,
, and
; Bottom side
,
,
,
,
, and
).
In this letter, the back-to-back E-shaped DGS and the twoside loading scheme of different-sized DGSs on top and bottom
sides of SIW are proposed for miniaturized dual-band ﬁlter designs. Based on the analysis and investigations, a 2.4/5.2 GHz
BPF was designed and veriﬁed experimentally. It is noted that
the proposed back-to-back E-shaped DGS for SIW and loading
scheme can be extended for miniaturized multi-band components.
REFERENCES
Fig. 6. Photographs, simulated and measured S-parameters of the proposed
ﬁlter.
element of the Chebyshev low-pass ﬁlter prototype [11]. The
example DGS ﬁlter operated in 5.2 GHz loaded on the bottom
side of SIW is given in Fig. 4.
III. DUAL-BAND BANDPASS FILTER PROTOTYPE
As shown in Fig. 5, a dual-band BPF is designed by loading
DGS resonators on both top and bottom sides. A back-to-back
E-shaped DGS ﬁlter on top side for the 5.2 GHz band and a
DGS on bottom side for the 2.4 GHz band are proposed. The
2.4 GHz ﬁlter is designed ﬁrst with fractional bandwidth (FBW)
of 5.8%. The coupling coefﬁcient is calculated as
and
. With 2.4 GHz resonator loaded on the bottom
side, the 5.2 GHz passband is designed on the top side with FBW
of 6.5%, the coupling matrix is generated as
and
. Since the frequency 5.2 GHz is 2.17 times of
that of the 2.4 GHz band, the size of the 5.2 GHz BB-E-shaped
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