Download Report

MIC285
511
60
0VIN, 3A Sy
ynchronous
s Buck Regulator
Gen
neral Desc
cription
Featu
ures
The MIC28511 is
i a synchro
onous step-d
down switching
regullator with inte
ernal power sw
witches capable of providing
up to
o 3A output current from a wide inpu
ut supply range
from 4.6V to 60V. The output voltage
v
is adju
ustable down to
0.8V with a gua
aranteed accuracy of ±1%
%. A consta
ant
switcching frequen
ncy can be prrogrammed from
f
200kHz to
680kkHz. The Hy
yper Speed Control™ and
a
HyperLig
ght
Load
d® architectures of the MIC28511 allo
ow for high VIN
(low VOUT) opera
ation and ultra-fast trans
sient response
while
e reducing the required
d output capacitance and
providing very goo
od light-load efficiency.
e




The MIC28511 offfers a full su
uite of protecttion features to
ensure protection
ns under fault conditions. These include
unde
er-voltage loc
ckout to ensu
ure proper operation
o
und
der
powe
er sag condittions, internal soft start to
o reduce inrush
curre
ent, foldback current limit, “hiccup” mo
ode short-circ
cuit
prote
ection and the
ermal shutdow
wn.
Datasheets and support
s
docu
umentation arre available on
Micre
el’s web site at:
a www.micre
el.com.








4.6V
V to 60V operrating input vo
oltage supply
Up tto 3A output ccurrent
Integ
grated high-sside and low-sside N-channe
el MOSFETs
Hyp
perLight Load (MIC28511-1) and H
Hyper Speed
d
Con
ntrol (MIC28511-2) architeccture
Ena
able input and
d power good (PGOOD) ou
utput
grammable current limit an
nd foldback ““hiccup” mode
e
Prog
shorrt-circuit prote
ection
Builtt-in 5V regula
ator for single-supply opera
ation
Adju
ustable 200kH
Hz to 680kHz switching fre
equency
Fixe
ed 5ms soft-sttart
Interrnal compenssation and the
ermal shutdow
wn.
The rmally-enhan
nced 24-pin 3m
mm × 4mm F
FCQFN
packkage
Juncction tempera
ature range off –40C to +125C
Appllications





Indu
ustrial power ssupplies
Disttributed supply regulation
wer supplies
Bas e station pow
Wal l transformer regulation
h-voltage sing
gle board systems
High
Typ
pical Application
Effficiency (VIN =12V
V)
vs. Outp
put Current MIC28
8511-1
100
5.0V
90
3.3V
EFFICIENCY (%)
80
2.5V
70
60
50
40
30
20
10
0.01
fSW = 300kHz
3
0.1
1
10
OU
UTPUT CURRENT (A
A)
Hype
er Speed Control and Ramp Control are trademarks of Micrel, Inc
c.
Hype
erLight Load is a registered trade
emark of Micrel, Inc.
I
Micrel Inc. • 2180 Fortune Driv
ve • San Jose, CA
C 95131 • USA • tel +1 (408) 94
44-0800 • fax + 1 (408) 474-1000
0 • http://www.m
micrel.com
March
h 25, 2015
Revision 1.2
Micre
el, Inc.
MIC28511
Ord
dering Info
ormation
Archite
ecture
Packa
age(1)
Junctio
on Temperaturre Range
Lead Finish
MIC
C28511-1YFL
HyperLight Load
24-Pin
2
3mm × 4mm FCQFN
–
–40°C to +125°°C
Pb-Free
MIC
C28511-2YFL
Hyper Spee
ed Control
24-Pin
2
3mm × 4mm FCQFN
–
–40°C to +125°°C
Pb-Free
Partt Number
Note:
1. FC
CQFN is a lead-ffree package. Pb
b-free lead finish is Matte Tin.
Pin Configuration
24-Pin 3mm
m × 4mm FCQF
FN (FL)
(Top View)
Pin Descriptiion
Pin
n Number
Pin Name
1
DL
2
PGND
3
DH
Pin Descriptio
on
Low-Side Gate
e Drive. Interna
al low-side pow
wer MOSFET ga
ate connection
n. This pin mustt be left
unconnected, or
o floating.
PGND is the re
eturn path for th
he low-side dri ver circuit. Con
nnect to the so
ource of low-sid
de MOSFET’s
(PGND, pins 10, 11 22, 23, and
a 26) through
h a low-impeda
ance path.
High-Side Gate
e Drive. Interna
al high-side pow
wer MOSFET gate connectio
on. This pin mu
ust be left
unconnected, or
o floating.
4, 7, 8, 9, 25
5 is ePad)
(25
PVIN
Power Input Vo
oltage. The PV
VIN pins supplyy power to the internal power switch. Connect all PVIN
pins together and
a bypass locally with ceram
mic capacitors. The positive te
erminal of the in
nput capacitor
should be plac
ced as close as
s possible to the
e PVIN pins, th
he negative terrminal of the inp
put capacitor
should be plac
ced as close as
s possible to the
e PGND pins 1
10,11, 22, 23, a
and 26.
5
LX
The LX pin is the return path for the high-sid
de driver circuit. Connect the negative termiinal of the
bootstrap capa
acitor directly to
o this pin. Also connect this p
pin to the SW pins 12, 21, and
d 27, with a
low-impedance
e path. The con
ntroller monitorrs voltages on tthis and PGND
D for zero curre
ent detection.
6
BST
March
h 25, 2015
Bootstrap Pin. This pin provid
des bootstrap ssupply for the h
high-side gate d
driver circuit. C
Connect a
0.1µF capacito
or and an optional resistor in sseries from the
e LX (pin 5) to tthe BST.
2
Revision 1.2
2
Micrel, Inc.
MIC28511
Pin Description (Continued)
Pin Number
Pin Name
10, 11, 22, 23,
26
(26 is ePad)
PGND
12, 21, 27
(27 is ePad)
SW
13
AGND
Analog Ground. The analog ground for VDD and the control circuitry. The analog ground return path
should be separate from the power ground (PGND) return path.
14
FB
Feedback Inout. The FB pin sets the regulated output voltage relative to the internal reference. This
pin is connected to a resistor divider from the regulated output such that the FB pin is at 0.8V when
the output is at the desired voltage.
15
PGOOD
The power good output is an open drain output requiring an external pull-up resistor to external bias.
This pin is a high impedance open circuit when the voltage at FB pin is higher than 90% of the
feedback reference voltage (typically 0.8V).
16
EN
Enable Input. The EN pin enables the regulator. When the pin is pulled below the threshold, the
regulator will shut-down to an ultra-low current state. A precise threshold voltage allows the pin to
operate as an accurate UVLO. Do not tie EN to VDD
17
VIN
Supply voltage for the internal LDO. The VIN operating voltage range is from 4.6V to 60V. A
ceramic capacitor from VIN to AGND is required for decoupling. The decoupling capacitor should be
placed as close as possible to the supply pin.
18
ILIM
Currrent Limit Setting. Connect a resistor from this pin to the SW pin node to allow for accurate
current limit sensing programming of the internal low-side power MOSFET.
19
VDD
Internal +5V Linear Regulator: VDD is the internal supply bus for the IC. Connect to an external
1µF bypass capacitor. When VIN is <5.5V, this regulator operates in drop-out mode. Connect VDD
to VIN.
20
PVDD
A 5V supply input for the low-side N-channel MOSFET driver circuit, which can be tied to VDD
externally. A 1μF ceramic capacitor from PVDD to PGND is recommended for decoupling.
24
FREQ
Switching Frequency Adjust pin. Connect this pin to VIN to operate at 680kHz. Place a resistor
divider network from VIN to the FREQ pin to program the switching frequency.
March 25, 2015
Pin Description
Power Ground. These pins are connected to the source of the low-side MOSFET. They are the
return path for the step-down regulator power stage and should be tied together. The negative
terminal of the input decoupling capacitor should be placed as close as possible to these pins.
Switch Node. The SW pins are the internal power switch outputs. These pins should be tied
together and connected to the output inductor.
3
Revision 1.2
Micrel, Inc.
MIC28511
Absolute Maximum Ratings(2)
Operating Ratings(3)
PVIN, VIN to PGND ........................................ 0.3V to 65V
VDD, PVDD to PGND ................................ ……0.3V to 6V
VBST to VSW, VLX ........ …………………..…………0.3V to 6V
VBST to PGND …………………..…………0.3V to (VIN +6V)
VSW, to PGND ... ………………………...-0.3V to (VIN +0.3V)
VLX, VFB, VPG, VFREQ, VILIM, VEN to AGND
……………………. .................... -0.3V to (VDD+ +0.3V)
PGND to AGND ………………......................-0.3V to +0.3V
Junction Temperature (TJ) ....................................... +150C
Storage Temperature (TS) ......................... 65C to 150C
Lead Temperature (soldering, 10s) ............................ 300C
ESD HBM Rating(4)...................................................... 1.5kV
ESD MM Rating(4) ......................................................... 150V
Supply Voltage (PVIN, VIN) .............................. 4.6V to 60V
Enable Input (VEN) ................................................. 0V to VIN
VSW, VFEQ, VILIM, VEN ....................................................................... 0V to VIN
Junction Temperature (TJ) ........................ 40C to 125C
Junction Thermal Resistance
3mm × 4mm FCQFN-24 (θJA) ............................ 30°C/W
Electrical Characteristics(5)
VIN = 12V; TA = 25°C, unless noted. Bold values indicate 40°C ≤ TJ ≤ +125°C.
Parameter
Condition
Min.
Typ.
Max.
Units
60
V
Power Supply Input
4.6
Input Voltage Range (PVIN, VIN)
Quiescent Supply Current
Shutdown Supply Current
VFB = 1.5V (MIC28511-1)
0.4
0.75
VFB = 1.5V (MIC28511-2)
0.7
1.5
SW = unconnected, VEN = 0V
0.1
10
µA
mA
VDD Supply
VDD Output Voltage
VIN = 7V to 60V, IVDD = 10mA
4.8
5.2
5.4
V
VDD UVLO Threshold
VVDD rising
3.8
4.2
4.6
V
VDD UVLO Hysteresis
400
Load Regulation @40mA
mV
0.6
2
4.0
0°C ≤ TJ ≤ 85°C (±1.0%)
0.792
0.8
0.808
40°C ≤ TJ ≤ 125°C (±2%)
0.784
0.8
0.816
5
500
%
Reference
Feedback Reference Voltage
FB Bias Current
VFB = 0.8V
V
nA
Enable Control
1.8
EN Logic Level High
V
0.6
EN Logic Level Low
EN Hysteresis
EN Bias Current
200
VEN = 12V
5
V
mV
40
µA
Notes:
2. Exceeding the absolute maximum ratings may damage the device.
3. The device is not guaranteed to function outside its operating ratings.
4. Devices are ESD sensitive. Handling precautions are recommended. Human body model, 1.5kΩ in series with 100pF.
5. Specification for packaged product only.
March 25, 2015
4
Revision 1.2
Micrel, Inc.
MIC28511
Electrical Characteristics(5) (Continued)
VIN = 12V; TA = 25°C, unless noted. Bold values indicate 40°C ≤ TJ ≤ +125°C.
Parameter
Condition
Min.
Typ.
Max.
VFREQ = VIN
450
680
800
Units
Oscillator
Switching Frequency
VFREQ = 50%VIN
340
Maximum Duty Cycle
Minimum Duty Cycle
85
VFB>0.8V
%
0
Minimum Off-time
110
200
kHz
%
270
ns
Internal MOSFETs
High-Side NMOS On-Resistance
51
m
Low-Side NMOS On-Resistance
28
m
Short-Circuit Protection
Current-Limit Threshold
VFB = 0.79V
30
14
0
mV
Short-Circuit Threshold
VFB = 0V
24
7
8
mV
Current-Limit Source Current
VFB = 0.79V
50
70
90
µA
Short-Circuit Source Current
VFB = 0V
25
36
43
µA
50
µA
95
%VOUT
Leakage
SW, BST Leakage Current
Power Good (PGOOD)
85
PGOOD Threshold Voltage
Sweep VFB from low-to-high
90
PGOOD Hysteresis
Sweep VFB from low-to-high
6
%VOUT
PGOOD Delay Time
Sweep VFB from low-to-high
100
µs
PGOOD Low Voltage
VFB < 90% × VNOM, IPGOOD = 1mA
70
TJ Rising
160
°C
15
°C
5
ms
200
mV
Thermal Protection
Overtemperature Shutdown
Overtemperature Shutdown
Hysteresis
Soft Start
Soft-Start Time
March 25, 2015
5
Revision 1.2
Micrel, Inc.
MIC28511
Typical Characteristics
VIN Shutdown Current
vs. Input Voltage
VIN Operating Supply Current
vs. Input Voltage MIC28511-1
VOUT = 5V
IOUT = 0A
fSW = 300kHz
1.6
1.2
0.8
0.4
5.8
40
5.6
VDD VOLTAGE (V)
SHUTDOWN CURRENT (µA)
30
20
10
5
10 15 20 25 30
35 40 45 50 55
10
15
Output Voltage
vs. Input Voltage MIC28511-1
20
25
30
35
40
45
50
55
3.4
3.3
3.2
3.1
VOUT = 5.0V
5
10
15
20
25
30
35
40
45
50
55
INPUT VOLTAGE (V)
Enable Threshold
vs. Input Voltage MIC28511-1
VDD UVLO Threshold
vs. Temperature MIC28511-1
60
5.0
RISING
ENABLE THRESHOLD (V)
3.5
IDD = 40mA
4.6
60
1.5
VIN = 4V TO 45V
VOUT = 3.3V
IOUT = 2A
4.8
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
3.6
5.0
4.0
5
60
5.2
4.2
0
0.0
IDD = 10mA
5.4
4.4
VEN = 0V
R16 = OPEN
1.2
FALLING
0.9
0.6
HYSTERESIS
0.3
VIN = 12V
IOUT = 0A
4.9
VDD THRESHOLD (V)
SUPPLY CURRENT (mA)
6.0
50
2.0
OUTPUT VOLTAGE (V)
VDD Voltage
vs. Input Voltage MIC28511-1
4.8
RISING
4.7
4.6
4.5
4.4
FALLING
4.3
4.2
4.1
4.0
0.0
3.0
0
5
10
15
20
25
30
35
40
5
45
10
15
30
35
40
45
50
55
-50
60
6
4
2
0
25
50
75
100
Feedback Voltage
vs. Temperature MIC28511-1
Enable Threshold
vs. Temperature MIC28511-1
125
1.7
VIN = 12V
VOUT = 5.0V
IOUT = 0A
0.808
ENABLE THRESHOLD (V)
8
-25
TEMPERATURE (°C)
0.812
VIN = 12V
VOUT = 5.0V
fSW = 300kHz
FEEBACK VOLTAGE (V)
CURRENT LIMIT (A)
10
25
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
Output Peak Current Limit
vs. Temperature MIC28511-1
20
0.804
0.800
0.796
VIN = 12V
VDD = 5V
1.6
1.5
1.4
RISING
1.3
1.2
1.1
1.0
FALLING
0.9
0
0.792
-50
-25
0
25
50
75
TEMPERATURE (°C)
March 25, 2015
100
125
0.8
-50
-25
0
25
50
75
TEMPERATURE (°C)
6
100
125
-50
-25
0
25
50
75
100
125
TEMPERATURE (°C)
Revision 1.2
Micrel, Inc.
MIC28511
Typical Characteristics (Continued)
Output Voltage
vs. Output Current MIC28511-1
Efficiency (VIN =12V)
vs. Output Current MIC28511-1
5.2
5.1
5.0
4.9
100
100
90
90
5.0V
80
80
3.3V
5.0V
70
3.3V
60
2.5V
EFFICIENCY (%)
VIN = 12V
VOUT = 5.0V
fSW = 300kHz
EFFICIENCY (%)
OUTPUT VOLTAGE (V)
5.3
Efficiency (VIN = 24V)
vs. Output Current MIC28511-1
50
40
2.5V
70
60
50
40
30
30
4.8
20
10
0.01
4.7
0.0
0.5
1.0
1.5
2.0
2.5
3.0
Switching Frequency
vs. Output Current MIC28511-1
2.5V
60
50
40
30
fSW = 300kHz
20
10
0.01
500
450
400
350
300
250
200
3.5
IC POWER DISSIPATION (W)
2.0
1.5
VIN =12V
fSW = 300kHz
TJMAX =125°C
JA = 30°C/W
0.0
40
55
70
85
AMBIENT TEMPERATURE (°C)
March 25, 2015
100
3.3V
0.2
0
0.5
1
1.5
2
2.5
OUTPUT CURRENT (A)
IC Power Dissipation
vs. Output Current MIC28511-1
IC Power Dissipation
vs. Output Current MIC28511-1
3
3.0
VIN ==24V
Vin
24V
fSW = 300kHz
1.5
1.0
5.0V
0.5
3.3V
2.5V
0.0
25
5.0V
OUTPUT CURRENT (A)
2.0
2.5V
5.0V
3.3V
0.4
2.5V
12V Input Thermal Derating
MIC28511-1
2.5
0.6
0.0
OUTPUT CURRENT (A)
3.0
0.8
1.0 1.2 1.4 1.6 1.8 2.0 2.2 2.4 2.6 2.8 3.0
10
10
VIN = 12V
fSW = 300kHz
1.0
150
IC POWER DISSIPATION (W)
1
1
1.2
VIN = 12V
VOUT = 5.0V
550
100
0.1
0.1
IC Power Dissipation
vs. Output Current MIC28511-1
IC POWER DISSIPATION (W)
SWITCHING FREQUENCY (kHz)
3.3V
70
fSW = 300kHz
OUTPUT CURRENT (A)
600
5.0V
OUTPUT CURRENT (A)
10
Efficiency (VIN = 48V)
vs. Output Current MIC28511-1
80
0.5
1
10
0.01
OUTPUT CURRENT (A)
90
1.0
20
OUTPUT CURRENT (A)
100
EFFICIENCY (%)
0.1
fSW = 300kHz
Vin
V
=24V
48V
IN =
fSW = 300kHz
2.5
2.0
1.5
5.0V
3.3V
1.0
2.5V
0.5
0.0
0
0.5
1
1.5
2
OUTPUT CURRENT (A)
7
2.5
3
0
0.5
1
1.5
2
2.5
3
OUTPUT CURRENT (A)
Revision 1.2
Micrel, Inc.
MIC28511
Typical Characteristics (Continued)
24V Input Thermal Derating
MIC28511-1
48V Input Thermal Derating
MIC28511-1
3.5
30
3.3V
2.5V
3.0
5.0V
3.3V
2.5V
2.5
2.0
1.5
VIN =24V
fSW = 300kHz
TJMAX =125°C
JA = 30°C/W
1.0
0.5
5.0V
2.5
2.0
1.5
VIN = 48V
fSW = 300kHz
TJMAX = 125°C
JA = 30°C/W
1.0
0.5
0.0
40
55
70
85
100
25
40
Output Voltage
vs. Input Voltage
70
85
5.1
5.0
VOUT = 5.0V
IOUT = 3A
4.8
4.7
5
20
25
30
35
40
45
50
10
15
55
30
35
40
45
50
55
60
15
VIN =12V
VOUT = 5.0V
FSW = 300kHz
12
0.804
0.800
0.796
9
6
3
0
-50
-25
0
25
50
75
100
125
-50
TEMPERATURE (°C)
INPUT VOLTAGE (V)
Output Voltage
vs. Output Current MIC28511-2
-25
0
25
50
75
100
125
TEMPERATURE (°C)
Efficiency (VIN = 12V)
vs. Output Current MIC28511-2
5.3
25
Output Peak Current Limit
vs. Temperature MIC28511-2
VIN = 12V
VOUT = 5.0V
IOUT = 0A
0.808
60
20
INPUT VOLTAGE (V)
0.792
15
VOUT = 5V
IOUT = 0A
fSW = 300kHz
6
100
CURRENT LIMIT (A)
FEEBACK VOLTAGE (V)
5.2
Efficiency (VIN = 24V)
vs. Output Current MIC28511-2
100
100
90
90
5.2
EFFICIENCY (%)
80
5.1
5.0
4.9
VIN = 12V
VOUT = 5.0V
fSW = 300kHz
4.8
0.5
1.0
1.5
2.0
OUTPUT CURRENT (A)
March 25, 2015
2.5
60
50
40
30
3.0
10
0.01
70
5.0V
3.3V
2.5V
60
50
40
30
fSW = 300kHz
20
4.7
0.0
80
5.0V
3.3V
2.5V
70
EFFICIENCY (%)
OUTPUT VOLTAGE (V)
55
0.812
10
12
Feedback Voltage
vs.Temperature MIC28511-2
5.3
5
18
AMBIENT TEMPERATURE (°C)
AMBIENT TEMPERATURE (°C)
4.9
24
0
0.0
25
OUTPUT VOLTAGE (V)
SUPPLY CURRENT (mA)
3.0
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
3.5
VIN Operating Supply Current
vs. Input Voltage MIC28511-2
0.1
1
OUTPUT CURRENT (A)
8
fSW = 300kHz
20
10
10
0.01
0.1
1
10
OUTPUT CURRENT (A)
Revision 1.2
Micrel, Inc.
MIC28511
Typical Characteristics (Continued)
Switching Frequency
vs. Output Current MIC28511-2
Efficiency (VIN = 48V)
vs. Output Current MIC28511-2
90
70
5.0V
3.3V
2.5V
60
50
40
30
fSW = 300kHz
20
0.1
1
400
350
300
250
200
VIN = 12V
150
0.8
0.6
0.4
5.0V
3.3V
0.2
2.5V
0.5
1.0
1.5
2.0
2.5
3.0
0.0
0.5
1.0
1.5
2.0
2.5
OUTPUT CURRENT (A)
12V Input Thermal Derating
MIC28511-2
IC Power Dissipation
vs. Output Current MIC28511-2
IC Power Dissipation
vs. Output Current MIC28511-2
3.0
2.5
3.3V
2.0
2.5V
1.5
VIN = 12V
fSW = 300kHz
TJMAX =125°C
JA = 30°C/W
Vin
V
=24V
24V
IN =
fSW = 300kHz
1.5
1.0
5.0V
3.3V
0.5
2.5V
55
70
85
100
VIN = 48V
fSW = 300kHz
2.5
2.0
1.5
5.0V
1.0
3.3V
2.5V
0.5
0.0
0.0
40
3.0
3.0
2.0
0.0
0.0
0.5
AMBIENT TEMPERATURE (°C)
1.0
1.5
2.0
2.5
3.0
0.0
0.5
1.0
1.5
2.0
2.5
3.0
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
24V Input Thermal Derating
MIC28511-2
48V Input Thermal Derating
MIC28511-2
3.5
3.5
5.0V
3.0
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
1.0
OUTPUT CURRENT (A)
5.0V
25
1.2
0.0
0.0
10
IC POWER DISSIPATION (W)
OUTPUT CURRENT (A)
450
VIN =12V
fSW = 300kHz
1.4
OUTPUT CURRENT (A)
3.5
0.5
1.6
500
100
10
0.01
1.0
1.8
550
IC POWER DISSIPATION (W)
EFFICIENCY (%)
80
600
IC POWER DISSIPATION (W)
SWITCHING FREQUENCY (kHz)
100
IC Power Dissipation
vs. Output Current MIC28511-2
2.5
3.3V
2.0
1.5
2.5V
VIN = 24V
fSW = 300kHz
TJMAX =125°C
JA = 30°C/W
1.0
0.5
5.0V
3.3V
2.5V
3.0
2.5
2.0
1.5
VIN = 48V
fSW = 300kHz
TJMAX = 125°C
JA = 30°C/W
1.0
0.5
0.0
0.0
25
40
55
70
85
AMBIENT TEMPERATURE (°C)
March 25, 2015
100
25
40
55
70
85
100
AMBIENT TEMPERATURE (°C)
9
Revision 1.2
Micre
el, Inc.
MIC28511
Fun
nctional Characteris
stics
March
h 25, 2015
10
Revision 1.2
2
Micre
el, Inc.
MIC28511
Fun
nctional Characteris
stics (Con
ntinued)
March
h 25, 2015
11
Revision 1.2
2
Micre
el, Inc.
MIC28511
Fun
nctional Characteris
stics (Con
ntinued)
March
h 25, 2015
12
Revision 1.2
2
Micre
el, Inc.
MIC28511
Fun
nctional Characteris
stics (Con
ntinued)
March
h 25, 2015
13
Revision 1.2
2
Micre
el, Inc.
MIC28511
Fun
nctional Diagram
March
h 25, 2015
14
Revision 1.2
2
Micre
el, Inc.
MIC28511
Fun
nctional Description
n
The MIC28511 is an adaptive on-time sync
chronous buc
ck
regullator with integrated high-side and
a
low-side
MOS
SFETs suitable for high-in
nput voltage to low-outpu
ut
voltage conversion applications. It is design
ned to operate
over a wide inputt voltage range (4.6V to 60V) which is
suitable for automotive and industrial application. The
outpu
ut is adjustab
ble with an ex
xternal resistive divider. An
adap
ptive on-time control schem
me is employed to produce
a con
nstant switching frequency
y in continuo
ous-conduction
mode
e and reduce
ed switching frequency
f
in discontinuous
d
sopera
ation mode
e, improving light-load
d efficiency
y.
Overrcurrent prote
ection is implemented by sensing low
wside MOSFET’s RDS(ON). The de
evice features internal softtstart,, enable, UVL
LO, and therm
mal shutdown..
It is n
not recomme
ended to use
e MIC28511 with an OFF
Fady-state
ope
time cclose to tOFF(M
during
ste
eration.
MIN)
The adaptive ON
N-time contrrol scheme results in a
consttant switchin
ng frequencyy in the MIC
C28511. The
e
actua
al ON-time a
and resulting
g switching ffrequency will
vary with the diffferent rising and falling times of the
e
extern
nal MOSFET
Ts. Also, the minimum tON results in a
lowerr switching fre
equency in hig
gh VIN to VOUTT applications
s.
Durin
ng load tran
nsients, the switching frequency is
chang
ged due to th e varying OF
FF-time.
Figurre 1 shows th
he allowable rrange of the o
output voltage
e
versu
us the input vvoltage. The minimum outtput voltage is
0.8V which is lim
mited by the
e reference voltage. The
e
maxim
mum output voltage is 24
4V which is llimited by the
e
intern
nal circuitry.
Theo
ory of Operattion
As illustrated in the Functio
onal Diagram
m, the outpu
ut
voltage of the MIC
C28511 is sensed by the feedback
f
(FB
B)
pin vvia voltage dividers R1 an
nd R2, and compared
c
to a
0.8V reference voltage
v
VREF at the erro
or comparato
or
throu
ugh a low-gain
n transconductance (gM) amplifier.
a
If the
feedb
back voltage decreases and
a
the amplifier output is
below
w 0.8V, then
n the error comparator
c
will
w trigger the
contrrol logic and generate an ON-time perriod. The ON
Ntime period length is predetermined by the fixed tON
O
estim
mator circuitry:
t ON(ESTIMATED ) 
VOUT
VIN  fSW
Eq. 1
wherre VOUT is the
e output volta
age, VIN is the
e power stage
inputt voltage, and fSW is the sw
witching freque
ency.
Figurre 1. Allowable
e Output Volta
age Range vs.. Input Voltage
e
At th
he end of the
e ON-time pe
eriod, the inte
ernal high-side
drive
er turns off th
he high-side MOSFET
M
and
d the low-side
drive
er turns on the
t
low-side MOSFET. The
T
OFF-time
perio
od length depe
ends upon the feedback voltage in mos
st
cases. When the
e feedback voltage decre
eases and the
ut of the gM amplifier is below 0.8V, then the ON-time
outpu
perio
od is triggered
d and the OF
FF-time perio
od ends. If the
OFF--time period determined by
b the feedba
ack voltage is
less than the minimum OFF-time tOFF(MIN), which
w
is abou
ut
200n
ns (typical), th
he MIC28511 control logic
c will apply the
tOFF(M
d to maintain
MIN) instead. The tOFF(MIN)) is required
enou
ugh energy in
n the boost ca
apacitor (CBST) to drive the
high--side MOSFE
ET.
To ill ustrate the ccontrol loop o
operation, botth the steady
ystate and load tran
nsient scenarrios will be analyzed.
Figurre 2 shows th
he MIC28511 control loop timing during
g
stead
dy-state ope
eration. Durin
ng steady-sttate, the gM
ampliifier senses the feedbackk voltage rip
pple, which is
propo
ortional to the
e output volta
age ripple and
d the inducto
or
curre nt ripple, to trrigger the ON
N-time period.. The ON-time
e
is pre
edetermined b
by the tON esttimator. The termination of
o
the O
OFF-time is co
ontrolled by th
he feedback vvoltage. At the
e
valleyy of the feedb
back voltage ripple, which
h occurs when
n
VFB fa
alls below VRREF, the OFF period ends and the nex
xt
ON-tiime period iis triggered through the control logic
circuiitry.
The m
maximum dutty cycle is obttained from:
D MAX  1  t OFF
O (MIN)  fSW
March
h 25, 2015
Eq. 2
15
Revision 1.2
2
Micre
el, Inc.
MIC28511
e true current-mode contrrol, the MIC28
8511 uses the
e
Unlike
outpu
ut voltage rip
pple to trigger an ON-time
e period. The
e
outpu
ut voltage riipple is proportional to the inducto
or
curre nt ripple if th
he ESR of the
e output capacitor is large
e
enoug
gh. The MIC
C28511 contro
ol loop has the advantage
e
of elim
minating the n
need for slope
e compensation.
In ord
der to meet th
he stability re
equirements, tthe MIC28511
feedb
back voltage ripple shou
uld be in ph
hase with the
e
inducctor current rip
pple and larg
ge enough to be sensed by
y
the gM amplifier and the error comparator. The
e
recom
mmended fee
edback voltage ripple is 20mV~100mV.
If a low-ESR ou
utput capacittor is selectted, then the
e
feedb
back voltage rripple may be
e too small to be sensed by
y
the g m amplifier a
and the erro
or comparatorr. Also, if the
e
ESR of the outp
put capacitorr is very low
w, the outpu
ut
voltag
ge ripple and
d the feedba
ack voltage rripple are no
ot
necesssarily in pha
ase with the inductor currrent ripple. In
n
these
e cases, ripple injection is required to e
ensure prope
er
opera
ation. Please refer to “Ripp
ple Injection” subsection in
n
Applic
ication Inform
mation for mo
ore details ab
bout the ripple
e
injecttion technique
e.
Figure 2.. MIC28511 Co
ontrol Loop Timing
Figurre 3 shows th
he operation of the MIC28
8511 during a
load transient. The
T
output voltage
v
drops
s due to the
sudden load incre
ease, which causes the VFB to be les
ss
than VREF. This will
w cause the error comparrator to trigge
er
an O
ON-time perio
od. At the end
d of the ON-ttime period, a
minim
mum OFF-tim
me tOFF(MIN) is generated to
o charge CBSST
since
e the feedbac
ck voltage is still below VREF. Then, the
next ON-time periiod is triggere
ed due to the low feedbac
ck
voltage. Thereforre, the switc
ching freque
ency change
es
durin
ng the load tra
ansient, but returns
r
to the nominal fixed
frequ
uency once th
he output has
s stabilized att the new load
curre
ent level. With the varying
g duty cycle and switching
frequ
uency, the outtput recovery
y time is fast and
a the outpu
ut
voltage deviation is small in MIC28511 conv
verter.
Disco
ontinuous M
Mode (MIC285
511-1 Only)
In co
ontinuous mode, the ind
ductor curre
ent is always
greatter than zero; however, at light loads th
he MIC285111 is able to forcce the inducctor current tto operate in
n
disco
ontinuous mo
ode. Discontin
nuous mode occurs when
n
the in
nductor curren
nt falls to zero
o, as indicate
ed by trace (IL)
show
wn in Figure 4. During thiis period, the
e efficiency is
s
optim
mized by shuttting down all the non-esssential circuits
and minimizing tthe supply ccurrent. The MIC28511-1
wake
es up and turn
ns on the hig
gh-side MOSF
FET when the
e
feedb
back voltage VFB drops bellow 0.8V.
The MIC28511-1 has a zero crossing comparator tha
at
monittors the inducctor current b
by sensing the
e voltage drop
p
acrosss the low-sid
de MOSFET during its O
ON-time. If the
e
VFB > 0.8V and the
e inductor currrent goes slig
ghtly negative
e,
then tthe MIC28511-1 automatically powers down most of
o
the IC
C circuitry and
d goes into a low-power m
mode.
511-1 goes into discontiinuous mode
Once
e the MIC285
e,
both DH and DL are low, which turns off the high-side
e
and l ow-side MOS
SFETs. The lload current iis supplied by
y
the o
output capacittors and VOUTT drops. If the
e drop of VOUT
cause
es VFB to go
o below VREFF, then all th
he circuits will
wake
e up into norrmal continuo
ous mode. F
First, the bias
curre nts of most circuitss reduced during the
e
disco
ed, and then a tON pulse is
ontinuous mod
de are restore
s
trigge
ered before tthe drivers arre turned on to avoid any
y
possiible glitches. Finally, the high-side drriver is turned
d
on. Figure 4 sshows the control loo
op timing in
n
disco
ontinuous mod
de.
Figure 3. MIC28511 Load Transient Re
esponse
March
h 25, 2015
16
Revision 1.2
2
Micre
el, Inc.
MIC28511
of the resistorr RILIM is comp
pared with the
e
The vvoltage drop o
low-sside MOSFET
T voltage dro
op to set the
e over-curren
nt
trip le
evel. The sma
all capacitor cconnected fro
om ILIM pin to
o
PGND
D can be added to filter tthe switching
g node ringing
g
allow
wing a betterr short limit measureme
ent. The time
e
consttant created b
by RLIM and th
he filter capaccitor should be
e
much
h less than the
e minimum offf time.
The overcurrent limit can be programm
med by using
g
Equa
ation 3:
R ILIM
M 
ICLIM  0.5  IL(PP)  R DS(ON)  VCL
ICL
Eq. 3
Wherre:
ICLIM = Desired currrent limit.
RDS(OON) = On-ressistance of llow-side pow
wer MOSFET
T
40mΩ
Ω (typical).
Figure 4. MIC28511-1
M
Control Loop Timing
T
(Discontinuous Mode)
VCL = Current-lim
mit threshold
d 14mV (typ
pical absolute
e
value
e). See the Ele
ectrical Chara
acteristics(5) ta
able.
Durin
ng discontinu
uous mode, the bias current of mos
st
circuits are reduc
ced. As a res
sult, the total power supply
curre
ent during dis
scontinuous mode
m
is only about 450μA,
A
allow
wing the MIC
C28511-1 to achieve high
h efficiency in
light load applicatiions.
ICL = Current-limit source curre
ent 70µA (typ
pical). See the
e
Electr
trical Characte
eristics(5) table
e.
∆IL(PPP) = Inductor ccurrent peak-to-peak (use Equation 4 to
o
calcu
ulate the inducctor ripple currrent).
VDD R
Regulator
The MIC28511 provides
p
a 5V
5 regulated VDD to bias
intern
nal circuitry fo
or VIN ranging from 5.5V to 60V. When
VIN iss less than 5.5V,
5
VDD sho
ould be tied to VIN pins to
bypa
ass the interna
al linear regulator.
The p
peak-to-peak inductor currrent ripple is:
IL(PP) 
Soft--Start
Soft-start reduces
s the powerr supply inrush current at
a
startu
up by controlling the outp
put voltage riise time while
the o
output capacittor charges.

Eq. 4
The MOSFET RDS(ON) varies 30% to
o 40% with
h
tempe
erature; therrefore, it is rrecommended to use the
e
RDS(OON) at max jun
nction temperrature with 20% margin to
o
calcu
ulate RILIM in E
Equation 3.
The M
MIC28511 im
mplements an internal digita
al soft-start by
ramp
ping up the 0.8V
0
referenc
ce voltage (VREF) from 0 to
100%
% in about 5m
ms with 9.7m
mV steps. This controls the
outpu
ut voltage ratte of rise at turn on, minimizing inrush
curre
ent and eliminating outputt voltage ove
ershoot. Once
the ssoft-start cycle ends, the related
r
circuittry is disabled
to red
duce current consumption.
In casse of hard sh
hort, the curre
ent limit thresshold is folded
d
down
n to allow an
n indefinite hard short o
on the outpu
ut
witho
out any destrructive effect.. It is manda
atory to make
e
sure that the indu
uctor current used to charrge the outpu
ut
capaccitor during ssoft start is under the folded short limitt;
otherw
rwise the supply will go in hiccup mode
e and may no
ot
be fin
nishing the so
oft start succe
essfully.
Currrent Limit
The MIC28511 us
ses the RDS(O
ernal low-side
ON) of the inte
powe
er MOSFET to
o sense over--current cond
ditions. In each
switcching cycle, the inducto
or current is
s sensed by
b
monitoring the low
w-side MOSF
FET during itts ON period
d.
The sensed volta
age, V(ILIM), is compared with
w the powe
er
groun
nd (PGND) affter a blanking
g time of 150ns.
March
h 25, 2015

VOUT  VIN(MAX
M )  VOUT
VIN(MAX )  fSW  L
Powe
er Good (PGOOD)
The p
power good (PGOOD) pin is an open
n drain outpu
ut
which
h indicates lo
ogic high whe
en the output is nominally
y
90% of its steady sstate voltage..
17
Revision 1.2
2
Micrel, Inc.
MIC28511
MOSFET Gate Drive
The Functional Diagram shows a bootstrap circuit,
consisting of DBST, CBST and RBST. This circuit supplies
energy to the high-side drive circuit. Capacitor CBST is
charged, while the low-side MOSFET is on, and the
voltage on the SW pin is approximately 0V. When the
high-side MOSFET driver is turned on, energy from CBST
is used to turn the MOSFET on. As the high-side
MOSFET turns on, the voltage on the SW pin increases
to approximately VIN. Diode DBST is reverse biased and
CBST floats high while continuing to bias the high-side
gate driver. The bias current of the high-side driver is less
than 10mA so a 0.1μF to 1μF is sufficient to hold the gate
voltage with minimal droop for the power stroke (highside switching) cycle, i.e. ∆BST = 10mA x 1.25μs/0.1μF =
125mV. When the low-side MOSFET is turned back on,
CBST is then recharged through the boost diode. A 30Ω
resistor RBST, which is in series with BST pin, is required
to slow down the turn-on time of the high-side N-channel
MOSFET.
March 25, 2015
18
Revision 1.2
Micre
el, Inc.
MIC28511
App
plication Informatio
on
Outp
put Voltage Setting
S
Comp
ponents
The MIC28511 re
equires two resistors
r
to set
s the outpu
ut
voltage as shown in Figure 5:
VIN
N
MIC28511
R19
9
FR
REQ
R17
7
Figure 5.
5 Voltage-Divider Configura
ation
Figure 6. S
Switching Freq
quency Adjus
stment
The o
output voltage
e is determine
ed by Equatio
on 5:
R1 

VOUT  VFB  1 

 R2 
Equa
ation 7 gives the estimated switching fre
equency:
Eq. 5
R17


FSW

W  F0  

R
19
R
7
17


Where: VFB = 0.8V
V
A typ
pical value of
o R1 used on
o the standa
ard evaluation
board
d is 10kΩ. If R1 is too larg
ge, it may allo
ow noise to be
introd
duced into th
he voltage fe
eedback loop
p. If R1 is too
small in value, it will
w decrease the efficiency
y of the powe
er
supp
ply, especially
y at light loads
s. Once R1 is
s selected, R2
can b
be calculated using Equation 6:
R2 
VFB
F  R1
VOU
UT  VFB
GND
Eq. 7
Wherre:
fO = Switching frrequency when R17 is o
open, 680kHz
z
typica
ally
Figurre 7 shows the switchin
ng frequencyy versus the
e
resisttor R17 when
n R19 = 100k
:
Eq. 6
Setting the Switc
ching Freque
ency
The MIC28511 sw
witching frequency can be adjusted by
b
changing the resistor
r
divid
der network
k from VIN
N.
Figure 7
7. Switching Frrequency vs. R17
March
h 25, 2015
19
Revision 1.2
2
Micrel, Inc.
MIC28511
Inductor Selection
Values for inductance, peak, and RMS currents are
required to select the output inductor. The input and
output voltages and the inductance value determine the
peak-to-peak inductor ripple current. Generally, higher
inductance values are used with higher input voltages.
Larger peak-to-peak ripple currents will increase the
power dissipation in the inductor and MOSFETs. Larger
output ripple currents will also require more output
capacitance to smooth out the larger ripple current.
Smaller peak-to-peak ripple currents require a larger
inductance value and therefore a larger and more
expensive inductor. A good compromise between size,
loss and cost is to set the inductor ripple current to be
equal to 20% of the maximum output current. The
inductance value is calculated by:
L

VOUT  VIN(MAX )  VOUT
VIN(MAX )  IL(PP)  fSW

The winding resistance must be minimized although this
usually comes at the expense of a larger inductor. The
power dissipated in the inductor is equal to the sum of the
core and copper losses. At higher output loads, the core
losses are usually insignificant and can be ignored. At
lower output currents, the core losses can be a significant
contributor. Core loss information is usually available
from the magnetics vendor. Copper loss in the inductor is
calculated by Equation 11:
PL(Cu) = IL(RMS)2 × DCR
The resistance of the copper wire, DCR, increases with
the temperature. The value of the winding resistance
used should be at the operating temperature.
Eq. 8
DCR(HT) = DCR20C × (1 + 0.0042 × (TH  T20C))
Where:
Where:
fSW = Switching frequency.
TH = Temperature of wire under full load.
L(PP) = The peak-to-peak inductor current ripple,
typically 20% of the maximum output current.
T20°C = Ambient temperature.
Output Capacitor Selection
The type of the output capacitor is usually determined by
its equivalent series resistance (ESR). Voltage and RMS
current capability are also important factors in selecting
an output capacitor. Recommended capacitor types are
ceramic, tantalum, low-ESR aluminum electrolytic, OSCON and POSCAP. For high ESR electrolytic capacitors,
ESR is the main cause of the output ripple. The output
capacitor ESR also affects the control loop from a
stability point of view. For a low ESR ceramic output
capacitor, ripple is dominated by the reactive impedance.
Eq. 9
The RMS inductor current is used to calculate the I2R
losses in the inductor.
IL(RMS)  I2 OUT(MAX ) 
I2L(PP)
I2
The maximum value of ESR is calculated:
Eq. 10
ESR COUT 
Maximizing efficiency requires the proper selection of
core material and minimizing the winding resistance. The
high frequency operation of the MIC28511 requires the
use of ferrite materials for all but the most cost sensitive
applications. Lower cost iron powder cores may be used
but the increase in core loss will reduce the efficiency of
the power supply. This is especially noticeable at low
output power. The winding resistance decreases
efficiency at the higher output current levels.
March 25, 2015
Eq. 12
DCR(20°C) = Room temperature winding resistance
(usually specified by the manufacturer).
In the continuous conduction mode, the peak inductor
current is equal to the average output current plus one
half of the peak-to-peak inductor current ripple.
IL (PK )  IOUT  0.5  IL(PP )
Eq. 11
VOUT(PP)
IL(PP)
Eq. 13
Where:
ΔVOUT(pp) = peak-to-peak output voltage ripple
∆IL(PP) = peak-to-peak inductor current ripple
20
Revision 1.2
Micrel, Inc.
MIC28511
The total output ripple is a combination of the ESR and
output capacitance. The total ripple is calculated by
Equation 14:
 2  IL(PP ) 
  IL (PP )  ESR COUT
VOUT (PP )  

 C OUT  f SW  8 

Input Capacitor Selection
The input capacitor for the power stage input VIN should
be selected for ripple current rating and voltage rating.
Tantalum input capacitors may fail when subjected to
high inrush currents, caused by turning the input supply
on. A tantalum input capacitor’s voltage rating should be
at least two times the maximum input voltage to
maximize reliability. Aluminum electrolytic, OS-CON, and
multilayer polymer film capacitors can handle the higher
inrush currents without voltage de-rating. The input
voltage ripple will primarily depend on the input
capacitor’s ESR. The peak input current is equal to the
peak inductor current, so:
2
Eq. 14
Where:
D = Duty cycle.
COUT = Output capacitance value.
VIN  IL(PK )  ESR CIN
fSW = Switching frequency.
The input capacitor must be rated for the input current
ripple. The RMS value of input capacitor current is
determined at the maximum output current. Assuming the
peak-to-peak inductor current ripple is low:
As described in the “Theory of Operation” section in the
Functional Characteristics section, the MIC28511
requires at least 20mV peak-to-peak ripple at the FB pin
for the gm amplifier and the error comparator to operate
properly. Also, the ripple on FB pin should be in phase
with the inductor current. Therefore, the output voltage
ripple caused by the output capacitors value should be
much smaller than the ripple caused by the output
capacitor ESR. If low-ESR capacitors, such as ceramic
capacitors, are selected as the output capacitors, a ripple
injection method should be applied to provide the enough
feedback voltage ripple. Please refer to the “Ripple
Injection” section for more details.
ICIN(RMS)  IOUT(MAX )  D  1  D 
IL(PP )
12
PDISS(CIN)  I2 CIN(RMS)  ESR CIN
March 25, 2015
Eq. 19
Ripple Injection
The VFB ripple required for proper operation of the
MIC28511’s gm amplifier and error comparator is 20mV to
100mV. However, the output voltage ripple is generally
designed as 1% to 2% of the output voltage. If the
feedback voltage ripple is so small that the gm amplifier
and error comparator can’t sense it, then the MIC28511
will lose control and the output voltage is not regulated. In
order to have some amount of VFB ripple, a ripple
injection method is applied for low output voltage ripple
applications.
Eq. 15
The power dissipated in the output capacitor is:
PDISS(COUT )  I2 COUT (RMS)  ESR COUT
Eq. 18
The power dissipated in the input capacitor is:
The voltage rating of the capacitor should be twice the
output voltage for a tantalum and 20% greater for
aluminum electrolytic or OS-CON. The output capacitor
RMS current is calculated by Equation 15:
ICOUT(RMS ) 
Eq. 17
Eq. 16
21
Revision 1.2
Micre
el, Inc.
MIC28511
The applications
s are divide
ed into three situations
according to the amount
a
of the feedback volltage ripple:
1. E
Enough ripple
e at the feedback voltag
ge due to the
la
arge ESR of the
t output capacitors.
A
As shown in Figure
F
8, the converter is stable withou
ut
a
any ripple inje
ection. The fee
edback voltag
ge ripple is:
VFB(PP) 
R2
E COUT  IL(PP)
 ESR
R1  R 2
Fig
gure 9. Inadeq
quate Ripple
Eq. 20
2
w
where ∆IL(pp) is
s the peak-to
o-peak value of
o the inducto
or
ccurrent ripple.
2. Inadequate rip
pple at the fe
eedback volta
age due to the
ssmall ESR of the output ca
apacitors.
T
The output voltage
v
ripple
e is fed into
o the FB pin
through a feed forward cap
pacitor CFF in this situation
n,
a
as shown in Figure 9. The typical CFF value is
sselected by:
Fiigure 10. Invis
sible Ripple
R1 CFF 
10
fSW
In thiis situation, the output vvoltage ripple
e is less than
n
20mV
V. Therefore, additional rip
pple is injecte
ed into the FB
B
pin frrom the switcching node SW
W via a resisstor RINJ and a
capaccitor CINJ, as shown in Fig
gure 10. The injected ripple
e
is:
Eq. 21
2
W
With the feed forward capa
acitor, the fee
edback voltage
rripple is very close
c
to the output
o
voltage
e ripple:
VFB(PP )  ESR COUT  IL(PP )
∆VFB(pp)  VIN  K div  D  (1- D) 
2
Eq. 22
3. V
Virtually no riipple at the FB pin voltag
ge due to the
vvery-low ESR
R of the outputt capacitors.
K DIV 
R1//R2
R IINJ  R1//R2
1
Eq. 23
fS
SW  
Eq. 24
Wherre:
VIN = Power stage
e input voltage
e
D=D
Duty cycle
fSW = Switching fre
equency
τ = (R
R1//R2//RINJ) × CFF
Figure
F
8. Enou
ugh Ripple
March
h 25, 2015
22
Revision 1.2
2
Micrel, Inc.
MIC28511
In Equations 23 and 25, it is assumed that the time
constant associated with CFF must be much greater than
the switching period:
1
fSW  

T
 1

Eq. 25
If the voltage divider resistors R1 and R2 are in the k
range, a CFF of 1nF to 100nF can easily satisfy the large
time constant requirements. Also, a 100nF injection
capacitor CINJ is used in order to be considered as short
for a wide range of the frequencies.
The process of sizing the ripple injection resistor and
capacitors is:
Step 1. Select CFF to feed all output ripples into the
feedback pin and make sure the large time constant
assumption is satisfied. Typical choice of CFF is 1nF to
100nF if R1 and R2 are in kΩ range.
Step 2. Select RINJ according to the expected feedback
voltage ripple using Equation 26:
K DIV 
∆VFB(pp)
VIN
f

 SW
D  (1 - D)
Eq. 26
Then the value of RINJ is obtained as:
R INJ  (R1//R2)  (
1
K DIV
 1)
Eq. 27
Step 3. Select CINJ as 100nF, which could be considered
as short for a wide range of the frequencies.
March 25, 2015
23
Revision 1.2
Micrel, Inc.
MIC28511
PCB Layout Guidelines
Input Capacitor
Warning: To minimize EMI and output noise, follow
these layout recommendations.
PCB Layout is critical to achieve reliable, stable and
efficient performance. A ground plane is required to
control EMI and minimize the inductance in power, signal
and return paths.
Figure 11 is optimized from small form factor point of
view shows top and bottom layer of a four-layer PCB. It is
recommended to use Mid-Layer 1 as a continuous
ground plane.

Place the input capacitors on the same side of the
board and as close to the PVIN and PGND pins as
possible.

Place several vias to the ground plane close to the
input capacitor ground terminal.

Use either X7R or X5R dielectric input capacitors. Do
not use Y5V or Z5U type capacitors.

Do not replace the ceramic input capacitor with any
other type of capacitor. Any type of capacitor can be
placed in parallel with the input capacitor.

If a Tantalum input capacitor is placed in parallel with
the input capacitor, it must be recommended for
switching regulator applications and the operating
voltage must be derated by 50%.

In “Hot-Plug” applications, a Tantalum or Electrolytic
bypass capacitor must be used to limit the overvoltage spike seen on the input supply with power is
suddenly applied.
SW Node

Do not route any digital lines underneath or close to
the SW node.

Keep the switch node (SW) away from the feedback
(FB) pin.
Output Capacitor
Figure 11. Top and Bottom Layer of a Four-Layer Board

The following guidelines should be followed to insure
proper operation of the MIC28511 converter:
Use a copper island to connect the output capacitor
ground terminal to the input capacitor ground
terminal.

Phase margin will change as the output capacitor
value and ESR changes. Contact the factory if the
output capacitor is different from what is shown in the
BOM.

The feedback trace should be separate from the
power trace and connected as close as possible to
the output capacitor. Sensing a long high-current load
trace can degrade the DC load regulation.
IC

The analog ground pin (AGND) must be connected
directly to the ground planes. Do not route the AGND
pin to the PGND pin on the top layer.

Place the IC close to the point of load (POL).

Use copper planes to route the input and output
power lines.

Analog and power grounds should be kept separate
and connected at only one location.
March 25, 2015
24
Revision 1.2
Micrel, Inc.
MIC28511
Thermal Measurements
Measuring the IC’s case temperature is recommended to
insure it is within its operating limits. Although this might
seem like a very elementary task, it is easy to get
erroneous results. The most common mistake is to use
the standard thermal couple that comes with a thermal
meter. This thermal couple wire gauge is large, typically
22 gauge, and behaves like a heatsink, resulting in a
lower case measurement.
Two methods of temperature measurement are using a
smaller thermal couple wire or an infrared thermometer. If
a thermal couple wire is used, it must be constructed of
36 gauge wire or higher then (smaller wire size) to
minimize the wire heat-sinking effect. In addition, the
thermal couple tip must be covered in either thermal
grease or thermal glue to make sure that the thermal
couple junction is making good contact with the case of
the IC. Omega brand thermal couple (5SC-TT-K-36-36) is
adequate for most applications.
Wherever possible, an infrared thermometer is
recommended. The measurement spot size of most
infrared thermometers is too large for an accurate
reading on a small form factor ICs. However, a IR
thermometer from Optris has a 1mm spot size, which
makes it a good choice for measuring the hottest point on
the case. An optional stand makes it easy to hold the
beam on the IC for long periods of time.
March 25, 2015
25
Revision 1.2
Micre
el, Inc.
MIC28511
MIC
C2851X Ev
valuation Board
B
Sch
hematic
Bill of Materials
Item
m
C1
C2, C3
C4, C7
Part Number
UVZ2A3
330MPD
12061Z4
475KAT2A
C1608X
X7R1A225K080
0AC
Man
nufacturer
Niichicon
(6)
C9
C10
0, C17
C0603C
C104K8RACTU
U
GRM21B
BR72A474KA7
73
08051C4
474KAT2A
GRM188
8R72A104KA3
35D
3
33µF/100V 20%
% Radial Aluminum Capacito
or
1
4
4.7µF/100V, X7
7S, Size 1206 Ceramic Capa
acitor
2
(8)
2
2.2µF/10V, X7R
R, Size 0603 C
Ceramic Capaccitor
2
O
OPEN
NA
0
0.1µF/10V, X7R
R, Size 0603 C
Ceramic Capaccitor
2
0
0.47µF/100V, X
X7R, Size 0805
5 Ceramic Cap
pacitor
1
0
0.1µF/100V, X7
7R, Size 0603 Ceramic Capa
acitor
2
AVX
TDK
(9)
Kemet
K
(9)
Murata
M
AVX
Murata
C11
C12
2
CGA3E2
2X7R1H471K
Qty.
(7)
C5, C13
C6, C16
D
Description
TDK
O
OPEN
NA
4
470pF/50V, X7
7R, Size 0603 C
Ceramic Capaccitor
1
Notes
s:
6. Niichicon: www.nic
chicon.co.jp/engliish.
7. AV
VX: www.avx.com
m.
8. TD
DK: www.tdk.com
m.
9. Ke
emet: www.keme
et.com.
10. Murata: www.mura
ata.com.
March
h 25, 2015
26
Revision 1.2
2
Micrel, Inc.
MIC28511
Bill of Materials (Continued)
Item
Part Number
C14, C15
GRM32ER71A476KE15L
Manufacturer
Murata
Description
Qty.
47µF/10V, X7R, Size 1210 Ceramic Capacitor
2
C18
Open
NA
C19
Open
NA
C20
Open
NA
C21
D1
C1608NP02A270J080AA
BAT46W-TP
TDK
(11)
MCC
D3
1
100V Small Signal Schottky Diode, SOD123
1
Open
J1, J7, J8, J10, J11,
J12, J16, J17, J18
77311-118-02LF
L1
XAL7030-682MED
R1
27pF 100V, NPO, Size 0603 Ceramic Capacitor
CRCW060310K0FKEA
FCI
(12)
Coilcraft(13)
Vishay Dale
(14)
NA
CONN HEADER 2POS VERT T/H
9
6.8µH, 10.7A sat current
1
10.0kΩ, Size 0603, 1% Resistor
1
R2
Open
NA
R9
Open
NA
R10
CRCW06033K24FKEA
Vishay Dale
3.24kΩ, Size 0603, 1% Resistor
1
R11
CRCW06031K91FKEA
Vishay Dale
1.91kΩ, Size 0603, 1% Resistor
1
R14, R15
CRCW06030000FKEA
Vishay Dale
0.0 Ω, Size 0603, Resistor Jumper
2
R26
R16, R17, R19, R3
Open
CRCW0603100K0FKEA
Vishay Dale
R25
100kΩ, Size 0603, 1% Resistor
Open
NA
4
NA
R18
CRCW06031K00JNEA
Vishay Dale
1.0kΩ, Size 0603, 5% Resistor
1
R20, R21
CRCW060349R9FKEA
Vishay Dale
49.9Ω, Size 0603, 1% Resistor
2
R22
CRCW06031K74FKEA
Vishay Dale
1.74kΩ, Size 0603, 1% Resistor
1
R23
CRCW08051R21FKEA
Vishay Dale
1.21Ω, Size 0805, 1% Resistor
1
R24
CRCW060310R0FKEA
Vishay Dale
10.0Ω, Size 0603, 1% Resistor
1
TP1  TP2
Open
TP7  TP14
77311-118-02LF
FCI
CONN HEADER 2POS VERT T/H
1
TP8  TP13
77311-118-02LF
FCI
CONN HEADER 2POS VERT T/H
1
TP17  TP18
77311-118-02LF
FCI
CONN HEADER 2POS VERT T/H
1
TP9, TP10, TP11,
TP12
1502
Keystone
(15)
Electronics
Testpoint Turret, .090
4
Micrel. Inc.(16)
60VIN, 3A Synchronous Buck Regulator
1
U1
MIC28511-1YFL
MIC28511-2YFL
Notes:
11. MCC: www.mccsemi.com.
12. FCI: www.fciconnect.com.
13. Coilcraft: www.coilcraft.com.
14. Vishay Dale: www.vishay.com.
15. Keystone Electronics: www.keystone.com.
16. Micrel Inc.: www.micrel.com.
March 25, 2015
27
Revision 1.2
Micrel, Inc.
MIC28511
MIC2851X Evaluation Board Layout
Top Layer
Mid Layer 1
March 25, 2015
28
Revision 1.2
Micrel, Inc.
MIC28511
MIC2851X Evaluation Board Layout (Continued)
Mid Layer 2
Bottom Layer
March 25, 2015
29
Revision 1.2
Micre
el, Inc.
MIC28511
Package Inforrmation and Recomm
mended Land Pattern
n(17)
24-P
Pin 3mm × 4mm
m FQFN Packa
age Type (FL))
Note:
17. Pa
ackage information is correct as of
o the publication
n date. For updattes and most currrent information , go to www.micrel.com.
March
h 25, 2015
30
Revision 1.2
2
Micrel, Inc.
MIC28511
MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA
TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com
Micrel, Inc. is a leading global manufacturer of IC solutions for the worldwide high-performance linear and power, LAN, and timing & communications
markets. The Company’s products include advanced mixed-signal, analog & power semiconductors; high-performance communication, clock
management, MEMs-based clock oscillators & crystal-less clock generators, Ethernet switches, and physical layer transceiver ICs. Company
customers include leading manufacturers of enterprise, consumer, industrial, mobile, telecommunications, automotive, and computer products.
Corporation headquarters and state-of-the-art wafer fabrication facilities are located in San Jose, CA, with regional sales and support offices and
advanced technology design centers situated throughout the Americas, Europe, and Asia. Additionally, the Company maintains an extensive network
of distributors and reps worldwide.
Micrel makes no representations or warranties with respect to the accuracy or completeness of the information furnished in this datasheet. This
information is not intended as a warranty and Micrel does not assume responsibility for its use. Micrel reserves the right to change circuitry,
specifications and descriptions at any time without notice. No license, whether express, implied, arising by estoppel or otherwise, to any intellectual
property rights is granted by this document. Except as provided in Micrel’s terms and conditions of sale for such products, Micrel assumes no liability
whatsoever, and Micrel disclaims any express or implied warranty relating to the sale and/or use of Micrel products including liability or warranties
relating to fitness for a particular purpose, merchantability, or infringement of any patent, copyright, or other intellectual property right.
Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product
can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical
implant into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A
Purchaser’s use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser’s own risk and Purchaser agrees to fully
indemnify Micrel for any damages resulting from such use or sale.
© 2014 Micrel, Incorporated.
March 25, 2015
31
Revision 1.2